Zero voltage switching half bridge resonant converter

ABSTRACT

A DC to DC converter combines a half-bridge inverter with a resonant rectifier through a series LC circuit which conducts power at substantially a single frequency. Energy stored in the parasitic capacitors of the two power switches are transferred from one parasitic capacitor to the other in order to enhance efficiency of operation. The transfer is controlled by controlling the relative phase between the voltage and current in the inverter section.

FIELD OF THE INVENTION

This invention is related to the field of resonant DC/DC power convertercircuits and more particularly to a high frequency resonant DC/DC powerconverter able to accept relatively high voltage input.

BACKGROUND OF THE INVENTION

The concepts of resonance and high frequency operation are beingincreasingly applied to the design of DC/DC power converter circuits.High frequency operation permits significant physical size and weightreduction of inductive and capacitive energy storage components. In oneparticular category of power supplies, referred to in one case asresonant converters and in another case as quasi-resonant converters,inductor-capacitor pairs operating at or near their resonance frequency,have been utilized to reduce switching losses in power switchingtransistor(s) by excluding the simultaneous presence of voltage andcurrent during switching transitions. In some converter s L-C pairs havebeen used to regulate output voltage and in that capacity have theadvantageous effect of reducing the bandwidth of energy transfer in theconverter

One approach to resonant DC to DC power conversion, disclosed in U.S.Pat. No. 4,607,323, combines a single ended inverter having zero voltageswitching transitions with a full wave rectifier. A matching networkcoupling the inverter to the rectifier controls the output impedance ofthe inverter to enable it to achieve the zero voltage switchingoperation so that switching loss in the switching transistor issignificantly reduced by substantially eliminating the simultaneousoccurrence of voltage and current in the switching transistor duringon-off and off-on switching transitions. However, the switchingtransistor is subjected to very high peak voltages during itsnonconduction interval in each cycle of operation that greatly exceedthe input voltage applied to it.

Another approach to achieving reduced switching loss in the switchingtransistor of a single ended power converter is disclosed in U.S. Pat.No. 4,415,959 which achieves high efficiency by operating with zerocurrent switching in the switching transistor and by substantiallyeliminating the simultaneous occurrence of voltage and current waveformstherein. This arrangement, however, limits the maximum frequency atwhich the converter may be operated because of losses which increase asa function of frequency and which are incurred by the discharge of thepower transistor's shunt capacitance. An alternative to this arrangementis the bridge type inverter which has a lower voltage stress across thepower switches. However a conventional bridge type such as a half-bridgeinverter cannot be efficiently used as an inverter for application in ahigh-frequency DC/DC converter at high frequency switched voltagesbecause energy is stored in the shunt capacitance of the switchingtransistors (normally FETs) and is then dissipated in the switchingtransistors.

In yet another approach, disclosed in the Apr. 1987 HFPC proceedings (B.Carsten--"A Hybrid Series-- Parallel Resonant Converter For HighFrequencies And Power Levels", pg. 41-47), low voltage stress of the FETswitches and zero voltage switching are achieved by using a half-bridgearrangement and controlling the switch off-time overlap.

Therefore the half bridge inverter as disclosed by Carsten isconstructed so as to be operative so that both switching transistors arecaused to be cyclically simultaneously nonconducting for a controlledperiod of time. This operational condition assures that energy stored inone transistor switch capacitance is transferred to the other transistorswitch capacitance without any appreciable energy dissipation in theswitch. The load presented to the inverter is inductive at the operatingfrequency. Hence, by controlling an interval of simultaneousnonconduction of the two switching transistors, a controlled currentflowing during the off time overlap is operative to discharge the energystored in one switching transistor capacitance into the other switchingtransistor capacitance.

A critical aspect of the high frequency converter is the rectificationprocess. The rectifying action of the rectifying diodes produces ringingsignals having high harmonics. The existence of parasitic elements inthe rectifier adds to this ringing signal generation. These highfrequency signals circulate throughout the rectifier and may causesignificant power loss. If the input impedance of the rectifier is tunedin a straight forward manner to assure a resistive input impedance, theinput resistance tracks the load resistance. If frequency shift controlis used as a regulation technique, that input resistance trackingcharacteristic requires a substantial bandwidth of frequency control fora given regulation range.

In the converter designs mentioned above, as well as in many similarpublished designs, while inverter switching loss has been significantlyreduced and switch voltage stress has been minimized, none of thedesigns has addressed the problem of taking the high-voltage,high-frequency output of the inverter and optimally transforming andrectifying it to obtain a low voltage DC output. Optimization in thissense means, in part, obtaining the desired range of input and outputregulation with a minimum of frequency shift in the inverter, as well asa minimum of dissipation loss in the inverter, and in the transformationand rectification components. These desired results are achieved by theproper selection of the rectification and transformation means and by amatching of the rectification and transformation means to the inverter.

BRIEF SUMMARY OF THE INVENTION

A DC/DC power converter embodying the principles of the inventioncombines a half-bridge resonant inverter with an efficient highfrequency resonant rectifier. The rectifier output voltage is regulatedby frequency control via a narrow band frequency controlled reactancewhich couples the inverter to the resonant rectifier.

The inverter section topology is based on the half bridge inverter, butis modified in operation to permit its efficient use at high frequency.In principle a half-bridge inverter alternately enables a DC powersource voltage and a ground reference voltage through a pair ofalternately switched power transistors to apply a square wave voltage toa series resonant circuit. The series resonant circuit presents asubstantially zero impedance to the fundamental of the switched squarewave voltage and hence the output of the inverter is a substantiallypure sinusoidal voltage waveform. Since the peak value of the squarewave voltage is no greater than the input voltage, a greater DC inputvoltage can be accommodated with a switching transistor having a givenpeak voltage capacity, than is possible in the case of single switchembodiments of the zero voltage switching type converter.

The half bridge inverter load in the illustrative embodiment of thisinvention includes a resonant rectifier which over the operatingfrequency range presents a linear impedance to the output of the halfbridge inverter. The resonant recitifier is operative in combinationwith circuit characteristic impedances implemented in the inverteritself as well as within the rectifier to control rectifying diodecurrent and voltage waveforms so that transient ringing responses aresubstantially eliminated. The input resistance of the rectifier also islinear in its waveform responses to input waveforms from the inverter.The resonant rectifier is also designed to provide a large voltagetransformation ratio thereby minimizing the requirement for magneticallyinduced high voltage transformation ratios and simplifying the magneticcomplexity. This resonant rectifier is further designed so that itsinput resistance is made to vary directly with power or in other wordsto vary inversely with rectifier load resistance for the purpose ofenhancing the regulation properties of the converter.

This enunciated novel circuit arrangement utilizing a zero voltageswitching half-bridge inverter in combination with a resonant rectifierwith high voltage transformation capability permits a high-frequencyDC/DC converter circuit to achieve substantial voltage level changesfrom input to output and to operate at high efficiency and accommodate ahigher input voltage for a given peak voltage rating of the switchingtransistor than is possible with prior art resonant DC/DC converters.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a regulated power supply embodying theprinciples of the invention,

FIG. 2 is a schematic of a half-bridge switched power train embodyingthe principles of the invention,

FIG. 3 is a schematic of a half-bridge switched power train embodyingthe principles of the invention and using power MOSFET devices as powerswitches,

FIG. 4 shows current and voltage waveforms to assist in describing theoperation of the power circuit shown in FIG. 3,

FIGS. 5-12 shows various stages of power switching and associatedwaveforms for a cycle of operation of the inverter in order to explainoperation of the half-bridge switching of the power train of FIG. 3,

FIG. 13 is a schematic of another half-bridge switched power trainembodying the principles of the invention, and

FIG. 14 is a schematic of a resonant rectifying arrangement suitable foruse in a half bridge converter embodying the principles of theinvention.

DETAILED DESCRIPTION

A block diagram of a high-frequency power converter embodying theprinciples of the invention is shown in FIG. 1. This converter of FIG. 1includes a half bridge inverter 110, which can accommodate a highvoltage input without subjecting its power switching devices to highvoltage stress greater than the input voltage. As described below thesepower switches are operated in a zero voltage switching mode in order toavoid dissipative losses at high operating frequencies due to energystorage in the inherent capacitances of the switching devices.

The output of the inverter 110 is coupled through a transformer 135 to aresonant rectifier 140 which is designed to provide rectificationwithout generating ringing diode waveforms and to provide a large LCimpedance transformation ratio and hence reduce the voltagetransformation ratio that needs to be supplied by the transformer 135.

A reactive series circuit 125 couples the inverter 110 to the resonantrectifier 140. It has relatively large sized inductive and capacitivereactances close in value to each other in order to permit regulation ofthe output voltage at terminal 195 via small changes in the switchingfrequency of the power switches of the inverter 110 under control ofvoltage regulation feedback circuitry.

While the converter circuit is shown as operating from a DC voltage, itis to be understood that this DC may be supplied by from line rectifiedAC as well as a DC voltage source.

This DC voltage is applied to the input terminal 101. In theillustrative example, the DC voltage may be considered to be quite high;(400 volts may be considered exemplary). A filter circuit 105 isprovided to protect the input line from harmonics generated by theswitching of the inverter. The inverter switching circuit 110 is ahalf-bridge type which can accommodate large input voltages. This halfbridge arrangement produces a square wave voltage at the circuit nodebetween the two power switches. The voltage across either of the twopower switches is limited to approximately the magnitude of the squarewave voltage plus small ringing transients. It inverts the applied DCvoltage and applies a controlled AC voltage signal to a series LCcircuit 125 which within the power switching frequency range is slightlyabove its resonance frequency. It is utilized in controlling the currentand voltage waveforms in and across the switching devices in theinverter switching circuit. This series LC circuit 125 is further usedto control the magnitude of power flow in order to regulate the outputvoltage. Such power flow regulation is controlled by varying thefrequency of operation of the inverter. An additional function of thisseries LC circuit includes the filtering of harmonics thereby assuringthat power transfer from the resonant circuit 125 to the transformer 135at the coupling crossed by line cut 127 occurs substantially at a singlefrequency, and constrains the current output of the resonant circuit tobe substantially sinusoidal. Furthermore the resonant circuit controlsthe phase between current and voltage waveforms in the inverter. Theoperation of the series LC circuit in controlling these signal waveformsand single frequency power flow is discussed below.

A transformer 135 couples the single frequency output of the resonantcircuit to a resonant rectifier 140 which is designed as discussed belowto have a controlled linear input impedance and to resonate diodeparasitic capacitances to minimize DC ripple voltage. The inductive andcapacitive elements of the rectifier also supply a high impedancetransformation ratio to augment the turns ratio voltage transformationsupplied by the transformer 135 permitting the use of a low turns ratiowhile achieving the desired DC voltage applied to output filter 150 andultimately appearing at output terminal 195.

One particular method of supplying drive to the switching devices of theinverter 110 is provided by a switch drive circuit 180 which may beembodied as a drive transformer. These switching devices are responsiveto regulation feedback circuitry to regulate the DC output voltage atoutput terminal 195. The sensed output voltage is connected, via lead158, to an error amplifier 160. The resulting error voltage, which isderived by comparison with a reference voltage 156 in error amplifier160, is applied to a voltage controlled oscillator 170, via an optoisolator 165. The frequency of the output of the voltage controlledoscillator 170 is determined by this error voltage. This oscillatoroutput then determines the frequency of operation of the inverter 110via the switch drive 180. Power flow through the series resonant circuit(which is at a single frequency) is regulated by adjusting the switchingfrequency of the inverter switching devices through the voltagecontrolled oscillator 170.

A schematic of a power converter embodying the invention is shown inFIG. 2 and shows a DC voltage source 200 (which may be rectified AC)connected to input terminals 201 and 202. Two power switches 211 and 221are shown connected in series; the series connection of those switchesbeing in shunt with the input DC voltage applied to input terminals 201and 202. Switches 211 and 221 are semiconductor power switches and areas shown in FIG. 3, subsequently, preferably MOSFET power devices. Eachswitch 211 and 221 preferably has an internal diode 212 and 222 shuntingits main power path and also has a parasitic capacitance 213 and 223also shunting its main power path. The parasitic capacitance of theswitches 211 and 221 is normally undesirable at high operatingfrequencies but the circuit is operated in a manner which permitsefficient circuit performance. Drive to the switches 211 and 221 isapplied to the drive terminals 281 and 282 respectively. Drive isarranged such that the two power switches 211 and 221 are alternatelyswitched to a conducting state with an intervening dead time and withthe phase between voltage and current controlled by the subsequentseries resonant circuit. The two power switches are each operated in azero voltage switching mode wherein neither switch is biased closeduntil the voltage across it has dropped to zero. By operating the powerswitches in a zero voltage switching mode, the energy stored in theinherent capacitance of the switch is not dissipatively dischargedthrough the switch.

The switched voltage output of the two power switches 211 and 221 isapplied at node 225 to a series resonant circuit comprising the inductor226 and the capacitor 227 connected in series. Their component valuesare chosen so that the resonant frequency of this series combination isslightly below the operating frequency of the converter so that itpresents an inductive impedance to the output of the power switches atnode 225. This LC impedance enables the two power switches 211 and 221to operate in a zero voltage switching mode by constraining the switchcurrent to lag the switch voltage and by setting the off time overlapbetween the two switches so as to discharge the switch capacitanceduring nonconduction. This LC circuit in addition to causing a switchcurrent lag with respect to voltage is also used to regulate power flowby operating as a voltage divider in series with the input impedance oftransformer 235. Varying the frequency of operation of the inverteralters the ratio of the impedance division and hence the size of thesignal at the input to the transformer 235, a primary winding 234 ofpower transformer 235. This transformer is shown with its magnetizinginductance 233 shunting its primary winding 234 which forms part of theload division network. A tuning capacitor 232 is shown connected inshunt with the primary winding 234.

The secondary winding 236 of transformer 235 is shown connected throughthe leakage reactance 237 to a diode 241 having a parasitic capacitance242 which resonates cooperatively with the leakage inductance 237 tocontrol the voltage wave shape across the diode 241 to be a non-ringingsingle pulse waveform. The inductive and capacitive elements of therectifier provide an impedance transformation which augments the turnsratio transformation of the transformer 235. Resonant rectifiers havebeen disclosed and discussed in detail in U.S. Pats. Nos. 4,449,174,4,605,333 and 4,684,041 whose teachings are incorporated by referenceinto this specification. These resonant rectifiers as described variablyin the references are characterized by: a constant input resistance overthe operating range of the converter as long as the DC load resistanceis constant; an input resistance that varies inversely with changes inthe DC load resistance; an input impedance defined by two frequencyresponse poles which bracket the operating frequency range of theconverter; and a linearity in the input impedance in which a drivingwaveform produces a similarly contoured resultant waveform. The outputof the recitifier is connected through an rf choke inductor 251 tooutput terminal 295. Lead 255 is connected to the error amplifier (notshown) of the regulation control. A load 299 to be energized isconnected to the output terminals 295 and 296.

A schematic of the power converter shown in FIG. 3 shows two powerMOSFET switching devices 311 and 321 connected in series with each otherand the series connection thereof in parallel with the input terminals301 and 302. A capacitor 303 shunts the input terminals 301 and 302 andfilters the input DC voltage which may be supplied by rectified ACvoltage. The power MOSFET switching devices 311 and 321 each include aparasitic capacitance 313 and 323 respectively shunting its mainconduction path. Internal body diodes 312 and 322 also shunt the mainconduction paths. Drive from the drive circuitry is applied to the twoMOSFET power switching devices through transformer 385. Drive signalsare applied to terminals 382 and 383 and to primary winding 384. Thesecondary windings 386 and 387 connected in opposing polarity areconnected across the gate-source terminals of the power MOSFET switchingdevices 311 and 321 respectively. The two MOSFET power switching devices311 and 321 are switched alternately with a controlled dead time betweenalternate conducting intervals when both power MOSFET switching devicesare nonconducting. Switching of these power MOSFET switching devices isat a frequency determined by a voltage controlled oscillator 120 in thefeedback circuit responsive to an error signal as shown in FIG. 1. Thedrive signals applied to the power MOSFET power switching devices 311and 321 are sinusoidal in waveform which signal waveforms are shown bywaveforms 410 and 420 in FIG. 4 respectively. The power MOSFET switchdevice is biased conducting when the amplitude of the applied sinusoidachieves the threshold drive levels 411 and 421 respectively. Thevoltage across each of the power MOSFET switching devices 311 and 321approximates a square waveform as shown by waveform 430 for power MOSFETswitch device 321. The switched output current at node 325 issubstantially sinusoidal as shown by current waveform 440 in FIG. 4 andis applied to a series LC circuit comprising the inductor 326 and thecapacitor 327 which constrains it to have this waveform. The converteris operated at a frequency which is slightly above the resonantfrequency of the series LC circuit. This enables the series LC circuitwhich when so operated has an inductive characteristic to control thephase between the current and voltage waveforms at the node 325 wherebythe sinusoidal current (waveform 440) lags the square wave voltage.Hence, during switching of the power MOSFET switching devices 311 and321 charge is swept out of the shunt parasitic capacitance of one powerMOSFET switching device and transferred to the other power MOSFETswitching device in order to substantially prevent switching losstherein.

By varying the frequency of switching the power output is regulated bythe frequency responsive impedance of the series LC circuit. Properselection of the parameter values of inductor 326 and capacitor 327permits a wide range of load regulation with a minimum of operatingfrequency variation. This is optimized with respect to the frequencybandwidth required for regulation over the expected load range when thereactance of each is a fairly high value. The current waveform appliedto the primary winding 334 of transformer 335 is further constrained bythe series resonant network to be substantially sinusoidal in nature.Hence the secondary winding 336 applies a substantially sinusoidalcurrent to the resonant rectifier which includes the rectifying diode341 and the rectifying diode 343.

The resonant rectifier is designed so that the parasitic capacitances342 and 344 of the rectifying diodes 341 and 343 resonate with theleakage inductances in response to the single frequency input in orderto eliminate high frequency ringing. Specific characteristics of thisrectifier as discussed in the above cited references and herein aboveinclude a substantially real input impedance that is inverselyresponsive to the output load impedance and an input impedance that issubstantially invariant to frequency variation in the normal frequencyoperational range. The resonant rectifier also provides impedancetransformation and hence the turns ratio that must be provided by thetransformer is significantly reduced. The input resistance is inverselyproportional to the DC output impedance and the real input impedance issubstantially constant over the operating range.

The operation and operating principles of the inverter section of theconverter may be readily understood by referring to the sequence ofFIGS. 5-12 which define and describe a typical cycle of operation of theinverter section of the converter in terms of eight specific timeintervals. It is important to keep in mind that while the two powerMOSFET switching devices are alternately switched with an interveningdead time the sinusoidal current flow through the series resonantcircuit connected to the node between the devices is continuous.

The initial state in the first time interval as shown in FIG. 5 haspower switch 511 closed (i.e. biased conducting) and switch 521 open.The current flow path is through a load network (not shown) connected toterminals 561 and 562 and current flow in the inverter is shown by thearrows i and has a sinusoidal waveform 501 for an interval exceeding 1/2of a cycle representing the current at node 525 entering the seriesresonant network (not shown) connected to terminal 561. The parasiticcapacitance 513 is in a discharged state and the parasitic capacitance523 has been fully charged to the input voltage level and hence the node525 is at the input voltage V_(o) as shown by voltage waveform 551.

The next time interval occurs when both power switches 611 and 621 areopen (nonconducting) as shown in FIG. 6. The parasitic capacitance 623discharges and the parasitic capacitance 613 accepts charge. The currentat node 625 continues to follow a sinusoidal waveform 602 however thevoltage at node 625 decreases to zero as shown by waveform 652. Thearrows 2 depicting current flow indicate the current is flowing throughboth parasitic capacitances 613 and 623.

By the third time interval shown in FIG. 7 parasitic capacitor 723 isfully discharged and body diode 722 becomes forward biased. The currentflow now is through the body diode 722 of switch 721 as shown by thecurrent arrows 3 and the current waveform 703 at node 725 is stillsinusoidal. The voltage level at node 725 is zero as shown by voltagewaveform 753.

The power switch 821 in FIG. 8 is closed at the beginning of the timeinterval four and the current flow, shown by sinusoidal current waveform804, is through the power switch 821 as shown by current arrows 4. Thevoltage at node 825 remains at zero as shown by voltage waveform 854.

Power switch 921 in FIG. 9 is still closed in the following fifth timeinterval but the direction of current flow has reversed as shown bycurrent arrows 5 and the negative sinusoidal current waveform 905. Thevoltage at node 925 remains at zero as shown by voltage waveform 955.

Both power switches are opened in the sixth time interval and thesinusoidal current shown by wave form 1006 in FIG. 10 follows thecurrent path shown by current arrows 6 which includes current flowthrough both parasitic capacitances 1013 and 1023. Capacitance 1013 isdischarged and capacitance 1023 is charged causing the voltage at node1025 to ramp up to the value of the input voltage as shown by voltagewaveform 1055.

Current conduction is through the body diode 1112 of switch 1111 in FIG.11 during the time interval seven as shown by the current arrows 7. Thecurrent at node 1125 still follows a sinusoidal waveform as shown bywaveform 1107. The voltage at node 1125 shown by waveform 1157 is at theinput voltage value.

The cycle of operation is completed in the time interval eight shown inFIG. 12. The switch 1211 is closed and the current shown by sinusoidalwaveform 1208 returns to zero. Current flow indicated by arrows 8 isthrough the power switch 1211. The voltage at node 1225 is at the inputvoltage level as shown by the voltage waveform 1258.

It is apparent from the foregoing description of the inverter operationand the accompanying current and voltage waveforms that the sinusoidalcurrent flow through the switch and its parasitic elements lags thesquare wave voltage appearing across the switches. This phase lagassures the transfer of charge from the parasitic capacitance associatedwith one switch to the parasitic capacitance associated with the otherswitch. This enhances the efficiency of operation of the inverter.

Another inverter topology is disclosed in FIG. 13 and includes two powerswitches 1311 and 1321 connected in a half bridge arrangement with inputvoltage source 1300. Each power switch includes inherent shuntcapacitance and shunt body diodes. Two balancing capacitors 1373 and1383 are included in the bridge inverter topology. The load and outputcircuits are connected between the midpoint nodes 1325 and 1375 of thepower switches and capacitive elements, respectively, and include atuned LC network comprising inductor 1326 and capacitor 1327 and anoutput transformer 1335, having it's primary winding 1334 in series withthe LC network. The secondary winding 1336 may be coupled to a rectifiercircuit (not shown).

This particular topology permits the transformer core to bebidirectionally driven thereby reducing the size requirements for agiven power rating. It is important, as described above, that thereactance of each of the inductor 1326 and capacitor 1327 be fairlylarge and close in value to each other to permit regulation within areasonably narrow frequency range.

A two diode resonant rectifier suitable for use in a converter embodyingthe principles of the invention is shown in FIG. 14. The substantiallysingle frequency input power signal to the rectifier is depicted by thesignal source 1400. An inductor 1411 and capacitor 1412 are connected inshunt across the rectifier input. Two rectifying diodes 1418 and 1419each include a parasitic capacitance designated as the shuntcapacitances 1416 and 1417 respectively. An inductor 1413 is connectedin series with the rectifying diode 1418, an inductor 1420 couples theoutput to the load 1425. The reactive elements including the shunt inputreactance, diode parasitic capacitor and the rectifier loop inductancemust all be tuned with respect to the operating frequency of theconverter to obtain two frequency response poles in the input impedancecharacteristic of the rectifier. With proper placement of theseimpedance poles on either side of the operating frequency range and witha properly loaded Q for these reactive elements, the input resistance ofthe rectifier becomes substantially independent of frequency within theoperating range of the converter. In addition the input resistancevaries in a manner which is inversely proportional to the DC loadresistance 1425.

What is claimed is:
 1. In combination:An inverter circuit, includinginput means for accepting a high voltage DC input; a resonant rectifierhaving a resistive component of input impedance that increases withincreasing power and including a significant reactive impedance forproviding a substantial impedance transformation and, connected torectify an output of the inverter circuit, the inverter including; afirst switch, a first capacitor shunting the first switch, a first diodeshunting the first switch, a second switch, a second capacitor shuntingthe second switch, a second diode shunting the second switch, the firstand second switch being connected in series with each other and acrossthe input means, a series LC circuit connected to couple a circuit nodebetween the first and second switches to the resonant rectifier, andhaving an inductive reactance over an operating frequency range of theinverter, and further having large inductive and capacitive reactiveimpedances close in value to each other so that a relatively smallfrequency change produces a large impedance variation, drive means forbiasing the first and second switches alternately conductive at aswitching frequency above a resonant frequency of the LC series circuit.2. The combination as defined in claim 1 whereinthe first and secondswitch are controlled to have common nonconducting intervals duringwhich charge is transferred from one of the first and second capacitorsto another one of the first and second capacitors.
 3. The combination asdefined in claim 1 wherein a resistive component of input impedance ofthe resonant rectifier is substantially invariant to changes infrequency within the operating range of the inverter.
 4. The combinationas defined in claim 1 and further including regulation circuitrycomprisingmeans for comparing an output voltage of the rectifier with areference voltage and generating an error voltage, and means for varyinga frequency of operation of the inverter in response to the errorvoltage.
 5. In combination:A switching circuit comprising; input meansfor accepting a DC voltage, a first switch and a second switch having acommon circuit and node conenction connected in a series connection andthe series connection connected across the input means, a first diodeand a first capacitor each shunting the first switch, a second diode anda second capacitor each shunting the second switch, an LC circuitcomprising inductance and capacitance connected in series and connectedto a circuit node common to the first and second switch, and inductiveand capacitive reactances having substantial values close in value toone another whereby a limited frequency change produces a substantialreactance change, an output network for coupling the tuned circuit to aload to be energized and including resonant rectification means which isadapted to present a substantially linear impedance to the tunedcircuit, and having sufficient inductive and capacitive reactance forproviding a substantial impedance transformation, means for alternatelydriving the first and second switch into conduction at a frequency abovea resonant frequency of the series LC circuit and providing controlledintervals of simultaneous nonconduction of the first and second switchbetween each alternate interval of conduction, each nonconductioninterval sufficient to allow a capacitor associated with one switch todischarge into a capacitor associated with another switch.
 6. Thecombination as defined in claim 5 and further includingmeans forregulating an output voltage applied to the load comprising a comparatorfor comparing the output voltage to a reference voltage and generatingan error voltage and means for controlling a frequency of the means foralternately driving in response to the error voltage.
 7. The combinationas defined in claim 5 whereby a resistive component at input impedanceof the rectification means increases with increasing power applied tothe load.
 8. The combination as defined in claim 5 wherein a currentoutput at the common switch node is constrained to be substantiallysinusoidal.
 9. A high frequency converter circuit comprising:a halfbridge configured inverter including first and second active switches,each including a shunt diode and a shunt capacitance, connected inseries and alternately enabled conducting for switching an applied DCvoltage with an interspersed controlled dead time between conductingintervals, a resonant rectifier including a rectifying diode andincluding sufficient associated reactive elements for producing asubstantial impedance transformation and operative for rectifying asubstantially single frequency signal without creating ringingtransients, a tuned reactive circuit including series connectedinductive and capacitive elements resonant at a frequency below aswitching frequency of the inverter and operative for coupling energyfrom a node between the first and second active switches of the inverterto the rectifier at substantially a single frequency and constraining acurrent waveform of an output of the inverter switches to lag thevoltage wave form across the first and second active switches, andfurther having substantial inductive and capacitive reactance close invalue to one another so that a slight frequency change produces asubstantial impedance change.
 10. A high frequency converter circuit asdefined in claim 9 wherein an input impedance of the resonant rectifieris substantially invariant to changes in frequency within the operatingrange of the converter.
 11. A high frequency converter circuit asdefined in claim 10 wherein a resistive component of an input impedanceof the resonant rectifier increases with increasing power applied to theload.
 12. In combination:an inverter circuit comprising; means foraccepting a high DC voltage input, a bridge type voltage switchingarrangement for inverting the high DC voltage input to a pulsed voltage,means for driving the bridge type switching arrangement within apreselected frequency range, a resonant rectifier circuit comprising;means for accepting a load to be energized, at least a rectifying diode,a rectifier input including a reactive shunt component operative forinsuring a resistive input impedance, the resonant rectifier circuithaving two operative frequency poles with resistive impedance maximalocated at opposite ends of a bandwidth of operation of the inverter andfurther having its resistive component of input impedance vary directlywith power applied to the rectifier input, a series LC circuit couplingthe bridge type voltage switching arrangement to the rectifier input andoperative to interact with the input impedance of the rectifier to forma variable voltage divider and being tuned to be resonant at a frequencybelow an operating frequency of the means for driving, means forgenerating an error signal in response to a deviation of a voltage atthe means for accepting a load from a regulated value, and meansresponsive to the error signal for varying an operating frequency of themeans for driving.
 13. A combination as claimed in claim 12 wherein theresonant rectifier includes reactive impedance operative for providing asubstantial impedance transformation of voltage applied to the rectifierinput.
 14. A DC to DC converter comprising:a half bridge inverter inwhich each power swtich includes a shunt diode and a shunt capacitance,the shunt diode being operative to prevent reverse voltages across eachpower switch, a resonant rectifier in which waveform shaping inhibitsringing across the rectifying diodes, and the reactive components definetwo frequency response poles related to the converter operatingfrequency so that an input impedance of the rectifier includes an inputresistance substantially independent of frequency within the operatingfrequency. a reactive circuit, inductive at the operating frequency,connected in series with the input impedance of the resonant rectifierand coupling each power switch of the half bridge inverter to theresonant rectifier, and voltage regulation circuitry operative forvarying a frequency of operation of the converter in order to vary animpedance of the reactive circuit and control a voltage applied to theinput impedance of the rectifier.
 15. A DC to DC converter as defined inclaim 14 wherein the reactive circuit comprises:a series LC circuithaving large inductive and capacitive reactive impedances close in valueto each other so that a relatively small frequency change produces alarge impedance variation.
 16. A DC to DC converter as defined in claim14 wherein the power switches are enabled alternately conducting with anintervening non-conduction interval sufficient to allow a shuntcapacitance associated with one power switch to discharge into a shuntcapacitance associated with another power switch.
 17. A DC to DCconverter as defined in claim 14 wherein the voltage regulationcircuitry includes:means for comparing an output voltage of the resonantrectifier with a reference voltage and generating an error voltage, andmeans for varying frequency of switching of the power switches inresponse to the error voltage.